Pilot aided adaptive minimum mean square interference cancellation and detection

ABSTRACT

A modified adaptive minimum mean square error (MMSE) receiver detects a spread spectrum signal, minimizing interference among multiple users. The receiver receives multiple pilot bits and corresponding data bits. A channel is estimated using at least one of the received pilot bits and, simultaneously, the data bits are detected based on the pilot bit and an inverse of a gain of the receiver. The spectrum signal may be spread in accordance with a wideband code division multiple access (W-CDMA) format, in which the pilot bits are modified to include a predetermined bit sequence of alternating ones and negative ones.

The present application is a continuation application of pending U.S.patent application Ser. No. 10/320,526 filed on Dec. 17, 2002, thesubject matter of which is expressly incorporated herein by reference inits entirety.

COPYRIGHT AUTHORIZATION

A portion of the disclosure of this patent document contains materialwhich is subject to copyright protection. The copyright owner has noobjection to the facsimile reproduction by anyone of the patent documentor the patent disclosure, as it appears in the Patent and TrademarkOffice patent file or records, but otherwise reserves all copyrightrights whatsoever.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to the field of telecommunications. Moreparticularly, the present invention relates to reducing interference incode division multiple access (CDMA) communication systems.

2. Acronyms

The written description provided herein contains acronyms which refer tovarious telecommunications services, components and techniques, as wellas features relating to the present invention. Although some of theseacronyms are known, use of these acronyms is not strictly standardizedin the art. For purposes of the written description, the acronyms aredefined as follows:

Additive White Gaussian Noise (AWGN)

Binary Phase Shift Key (BPSK)

Code Division Multiple Access (CDMA)

Direct Sequence Code Division Multiple Access (DS-CDMA)

Inter-Symbol Interference (ISI)

Maximum Likelihood (ML)

Minimum Mean Square Error (MMSE)

Multi-Access Interference (MAI)

Multi-User Detection (MUD)

Single User Detection (SUD)

Orthogonal Variable Spreading Factor (OVSF)

Quaternary Phase Shift Key (QPSK)

Universal Mobile Telecommunications Systems (UMTS)

Wideband Code Division Multiple Access (W-CDMA)

Wide Sense Stationary Uncorrelated Scattering (WSSUS)

3. Background Information

The communications industry has experienced significant growth in thedemand for wireless communications. The increased demand is related, inpart, to the improved quality and reliability of wireless networks,including mobile cellular networks, which have essentially evolvedthrough three generations. The first generation included analog systemsthat modulated voice signals onto radio frequency (RF) carrier waves,which were transmitted and received between base stations and mobileunits. The second generation of cellular networks introduced digitalencoding of analog voice signals, and included the time divisionmultiple access (TDMA) and code division multiple access (CDMA) cellularsystems. The second generation required a symmetric, full-duplex networkand was directed to accommodation of voice traffic. The third generationof cellular systems includes packet-switched transmissions and canaccommodate voice, data, audio and video communications.

With the increase in interest in commercial CDMA based mobile systems,as well as CDMA's dominance in third generation systems, research anddevelopment efforts have been directed to exploring receiver structuresand detection techniques. CDMA systems belong to the genre commonlyknown as spread spectrum systems, the use of which is wide spreadthroughout wireless telecommunications. In spread spectrum systems, datafrom the transmitted signal is spread over a predetermined bandwidth,such that the transmission bandwidth of each transmitted signal is muchhigher than the actual data bandwidth. Therefore, numerous CDMA signals,originating from different users, are essentially spread and multiplexedonto the same transmission bandwidth. The signals corresponding to thedifferent users are distinguishable by the CDMA receiver (or detector)based on the respective spreading codes or signatures.

CDMA systems include direct sequence CDMA (DS-CDMA), or direct sequencespread spectrum (DSSS), and frequency hopping CDMA. DS-CDMA, inparticular, codes transmitted signals in sequential channels, combiningthe transmitted bits with a higher sampling rate chip sequence andspreading the signal according to a spreading ratio. Because each bitis, in essence, redundantly represented by multiple chips, a spreadDS-CDMA data signal is more likely to be recovered by the receiver, evenwhen portions of the data are lost or encounter interference duringtransmission. DS-CDMA includes various conventional coding schemes, suchas IS-95, cdma2000 and wideband code division multiple access (W-CDMA).

The capacity of operational DS-CDMA systems is limited bycross-interference among the different user signals, all of which occupythe same frequency at the same time. The cross-interference amongsignals from different users is referred to as multi-access interference(MAI). Attempts to improve the capacity of CDMA systems includedreducing the level of perceived interference among the users, includingminimum mean square error (MMSE) detection. However, implementation ofan adaptive MMSE detection scheme in universal mobile telecommunicationssystems (UMTS) requires major changes in the physical layer and the datalink layer (layers 1 and 2, respectively) in the WCDMA air interface.Although a blind MMSE detection schemes (i.e., without trainingsequences) require only minor changes to W-CDMA air interface, they donot promise the kind of performance improvement as adaptive MMSE.

The receivers implemented in the second generation of CDMA systems wererelatively simple matched filter based (correlation detectors) singleuser receivers. By the early 1990s, it became apparent that CDMA systemsare not multi-access interference (MAI) limited. Rather, MAI was alimitation of the single-user Rake architecture incorporated into CDMAreceivers. Therefore, the combined CDMA and Rake systems needed to bechanged, or replaced, to minimize the MAI, as well as to account forinternal signal interference (ISI), caused by other data bitstransmitted from the same user.

Accordingly, the concept of multi user detection (MUD) was developed, bywhich a receiver jointly estimates all received signals from multipletransmission sources. However, an optimal MUD receiver, which is basedon joint maximum likelihood (ML) estimation of all users, is impracticalto implement because of the large computational complexity. MUDreceivers are complex in CDMA systems because they must appropriatelyestimate the channels of all the users, as compared to a traditionalsingle user detector, having the Rake architecture, which need onlyestimate a channel of a single user.

In a traditional CDMA system, the estimated parameters of the channelare h_(k)(t) and τ_(k), the amplitude/phase variation and the delayassociated with each path, respectively. In a maximal ratio combiningprocess, the receiver compensates for the delay and the amplitude/phasevariation to obtain the maximum possible signal to noise ratio. Aninherent limitation of the Rake architecture is its inability tosuppress the residual cross-correlation between signals with differentsignatures.

The present invention overcomes the problems associated with the priorart, as described below.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is further described in the detailed descriptionthat follows, by reference to the noted drawings by way of non-limitingexamples of embodiments of the present invention, in which likereference numerals represent similar parts throughout several views ofthe drawings, and in which:

FIG. 1 is a block diagram showing an exemplary packet-based wirelesstelecommunications network, according to an aspect of the presentinvention;

FIG. 2 is a block diagram showing an exemplary adaptive MMSE receiver,according to an aspect of the present invention;

FIG. 3 is a simplified depiction of an exemplary CDMA signal, accordingto an aspect of the present invention;

FIG. 4 is a simplified depiction of a received signal construction,according to an aspect of the present invention; and

FIG. 5 is a flowchart of exemplary application logic for detectingspread spectrum signals at the exemplary MMSE receiver, according to anaspect of the present invention.

BRIEF DESCRIPTION OF THE APPENDICES

Appendix A is a general background description of MMSE receivers,relating to aspects of the present invention; and

Appendix B is a general background description of CDMA receivers withrespect to time varying channel models, relating to aspects of thepresent invention.

DETAILED DESCRIPTION OF EMBODIMENTS

The present invention relates to a pilot aided, adaptive MMSE detectionscheme with pilot aided joint channel estimation and data bit detection.It includes the existing features of a W-CDMA air interface, along withsignal processing techniques to implement an adaptive MMSE detectionscheme in a UMTS network. The adaptive MMSE architecture of the presentinvention requires minimal changes in the W-CDMA air interface andprovides performance gains in the range of traditional MMSEarchitectures. Also, the MMSE detection scheme does not involve thetraditional type of CDMA receiver, e.g., including Rake receiving andfiltering, and improves the capacity of UMTS communication technologyfor voice and data.

The pilot aided adaptive MMSE detection receiver of the presentinvention minimally impacts W-CDMA protocol, yet improves capacitycomparable to that of a traditional MMSE detection receiver. Theadaptive MMSE detection receiver uses the common and/or dedicated pilotchannels (already present in W-CDMA) to jointly estimate the channel andthe signal in a single step. The underlying detection algorithm reducesthe level of interference significantly, as compared to a traditionalW-CDMA receiver, and accordingly improves the capacity of the network.

In view of the above, the present invention through one or more of itsvarious aspects and/or embodiments is presented to accomplish one ormore objectives and advantages, such as those noted below.

An aspect of the present invention provides a method detecting a spreadspectrum signal, including pilot bits and corresponding data bits, at anadaptive MMSE error receiver. The method includes receiving the pilotbits and the corresponding data bits, estimating a channel using atleast one of the pilot bits, and simultaneously detecting the data bitsbased on the pilot bit and an inverse of a gain of the receiver. Thedata bits may be spread in accordance with a W-CDMA format, while thepilot bits may be spread in accordance with a modified W-CDMA format,which includes a predetermined bit sequence of alternating ones andnegative ones.

Simultaneously detecting the data bits may further include determining aphase reference with respect to the at least one pilot bit. Also,simultaneously detecting the data bits may be further based on a databit estimation vector and a pilot bit estimation vector determinedthough sampling a predetermined number of received signals.

Another aspect of the present invention provides a method for detectingspread spectrum signals at an adaptive MMSE receiver, in which spreadspectrum signals are received, including a pilot signal portion and adata signal portion. The pilot signal portion is estimated based on themultiple received spread signals and at least one pilot signal spreadingsequence corresponding to the pilot signal portion. The data signalportion is estimated based on the received multiple spread signals andat least one data signal spreading sequence corresponding to the datasignal portion. A data bit of the data signal portion is detected basedon the estimated pilot signal portion, the estimated data signal portionand at least one pilot bit of the pilot signal portion. The pilot signalportion may be a common pilot signal or a dedicated pilot signalcorresponding to the detected data bit.

Another aspect of the present invention provides a system for detectinga spread spectrum signal received at an adaptive MMSE receiver,including a demodulator, a channel estimator and a bit estimator. Thespread spectrum signal includes multiple pilot bits and correspondingdata bits. The demodulator demodulates the received pilot bits and databits. The channel estimator estimates a channel using at least one ofthe pilot bits. The bit estimator detects the data bits, concurrentlywith the channel estimator estimating the channel, based on the at leastone pilot bit and an inverse of a gain of the receiver. The bitestimator further base detection on an expectation value of the receivedsignal, determined though sampling a predetermined number of receivedsignals.

Another aspect of the present invention provides a system for detectingtransmitted spread spectrum data at an adaptive MMSE receiver, includinga demodulator, which receives and demodulates multiple received spreadspectrum signals, and a covariance matrix calculator, which determines areceived signal expectation value based on the received signals. Thesystem further includes a channel estimator, which determines a pilotspreading sequence of the received signals and estimates at least onepilot portion of the received signals based on the pilot spreadingsequence, and the expectation value, and a bit estimator whichdetermines a bit spreading sequence of the received signals andestimates at least one data portion of the received signals based on thebit spreading sequence and the expectation value. A combiner combinesthe at least one pilot portion and the at least one data portion andestimates the transmitted spread spectrum data based on the combinedpilot portion and data portion and a phase reference between the dataportion and at least one pilot bit of the pilot portion. The at leastone pilot bit may be a common pilot signal or a dedicated pilot signal,corresponding to the detected data.

Another aspect of the present invention provides computer readablemedium, storing a computer program for detecting a spread spectrumsignal at an adaptive MMSE receiver. The program includes a receivingsource code segment, a channel estimating source code segment and adetecting source code segment. The receiving source code segment thatreceives pilot bits and corresponding plurality of data bits in thespread spectrum signal. The channel estimating source code segmentestimates a channel using at least one of the pilot bits. The detectingsource code segment detects the data bits, contemporaneously with thechannel estimating source code segment estimating the at least one ofpilot bit, based on the pilot bit and an inverse of a gain of thereceiver. The detecting source code segment may contemporaneously detectthe bits further based on a phase reference with respect to the at leastone pilot bit or based on a data bit estimation vector and a pilot bitestimation vector determined though sampling a predetermined number ofreceived signals.

Yet another aspect of the present invention provides a computer readablemedium, storing a computer program for detecting spread spectrum signalsat an adaptive MMSE receiver. A receiving source code segment of thecomputer program receives multiple spread signals, which include a pilotsignal portion and a data signal portion. A pilot signal estimatingsource code segment estimates the pilot signal portion based on thespread signals and at least one pilot signal spreading sequencecorresponding to the pilot signal portion. A data signal estimatingsource code segment estimates the data signal portion based on thereceived spread signals and at least one data signal spreading sequencecorresponding to the data signal portion. A detecting source codesegment detects a data bit of the data signal portion based on theestimated pilot signal portion, the estimated data signal portion and atleast one pilot bit of the pilot signal portion.

The various aspects and embodiments of the present invention aredescribed in detail below.

The present invention is implementable in a wireless network, includingthe second generation and third generation cellular systems, such as theexemplary wireless network 10 of FIG. 1, which is implemented accordingto well known cellular communication techniques. The wireless network 10includes cell A, cell B and cell C, located adjacent to one other. It isunderstood that, although only three cells are depicted, the number ofcells in an actual wireless network is usually much larger. Cell Aincludes a base station 12, configured to enable communications withmultiple wireless mobile stations, including for example mobile stations14 and 16, which are shown as located within cell A. Cells B and Clikewise include respective base stations (not pictured) that operatesubstantially the same as the base station 12. The mobile stations 14and 16 may include wireless communication devices, such as cellulartelephones, multi-media communicators, personal digital assistants(PDA), and the like, as well as devices such as desktop personalcomputers, laptop personal computers, facsimile machines, and the like,working through a cellular modem. Mobile stations 14 and 16 may movefrom cell to cell during the course of an active communication session,in which case cell A performs a coordinated handover or handoff to oneof the adjacent cells, whenever one of the mobile stations travels intocell B or cell C.

The depicted base station 12 and mobile stations 14 and 16 havecompatible transmitters and receivers to enable communications withinthe wireless network 10. The receivers are adaptive MMSE receivers thatdetect slightly modified CDMA signals in accordance with the presentinvention, a simplified example of which is depicted as receiver 200 inFIG. 2.

Referring to FIG. 2, a transmitted data signal is spread on a radiofrequency (RF) carrier and received at the antenna 201. The data signalpasses through a band pass receive filter 202 and a multiplexer 204 toremove the RF carrier, indicated by e^(−jωt), and to down convert thereceived signal from the pass band to a base band. The base band signalpasses through an analog to digital (A/D) converter 206, such as amatched filter, to obtain the digital data of the originally transmittedsignal. The digital data enters an adaptive MMSE algorithm processor209, which includes a covariance matrix calculator 208, a channelestimator 210, a bit estimator 212, dividers 220-222 and a summationblock 230. The functions of the adaptive MMSE algorithm processor 209,discussed in detail below, may be derived, for example, assumingdiscrete linear time varying channels, in an embodiment of the presentinvention. The receiver 200 outputs the detected data 232.

As stated above, the receiver 200 detects DS-CDMA data, having amodified pilot signal. FIG. 3 illustrates a simplified, exemplaryrepresentation of a typical DS-CDMA signal transmitted for a singleuser. General background information regarding DS-CDMA theory isprovided in Appendix B, the contents of which are incorporated byreference herein in its entirety. Each symbol or bit b is represented byseries of chips c. The number of chips c in each bit b depends on thespreading configuration of the transmitter. Typically, a bit b isrepresented by a multiple of two chips, such as 16, 32 or 64 chips perbit.

As shown in FIG. 3, the bit sequence 302 for user j is represented byb_(j)(i) and the chip sequence 303 (e.g., the spreading filterfunction), is represented by c_(j)(n), where i is the bit index and n isthe chip index within a given bit. T_(b) and T_(c) represent the bitduration and the chip duration in units of time, respectively, whereT_(b) is equal to T_(c) multiplied by the processing gain N (i.e.,T_(b)=N T_(c)). Both sequences are assumed to be binary, although inmany conventional DS-CDMA systems, the sequences are not binary due tohigher forms of modulation and complex spreading processes. Thesimplistic representation of the DS-CDMA signal in FIG. 3 aids indeveloping basic algebraic relationships incorporated into the adaptiveMMSE algorithm processor 209.

The bit sequence b_(j)(i) is multiplied by the chip sequence c_(j)(n) at305 to create a combined data sequence, represented asS_(j)(n)=b_(j)(i)c_(j)(n). The spreading of the data sequence isrepresented by a pulse shaping filter 307, which is user specific.Accordingly, the continuous, transmitted signal s at time t isrepresented as: $\begin{matrix}{{s(t)} = {\sum\limits_{i = {- \infty}}^{+ \infty}\quad{{b\lbrack i\rbrack}{\sum\limits_{n = 0}^{N - 1}\quad{c_{n + {iN}}{f\left( {t - {iT}_{s} - {nT}_{c}} \right)}}}}}} & (1)\end{matrix}$where the spreading filter can be represented by$\sum\limits_{n = 0}^{N - 1}\quad{c_{n + {iN}}{{f\left( {t - {iT}_{s} - {nT}_{c}} \right)}.}}$

Equation (1) does not assume that the spreading function is stationary.In other words, equation (1) does not assume that the chip sequencec_(j)(n) is the same for every bit b_(j), which is the case in manyDS-CDMA systems in which the orthogonal spreading codes, such asHadamard and orthogonal variable spreading factor (OVSF) codes, aremultiplied by long scrambling codes (i.e., PN sequences), such as IS-95,cdma2000 and W-CDMA. For example, the PN sequences for IS-95 andcdma2000 are ML shift register sequences and for W-CDMA are Gold codes.The dependence of the chip sequence c_(j)(n) on the bit index i makesthe spreading filter non-stationary. However, in practical CDMA systems,the spreading sequence repeats after a fixed interval, making thespreading filter cyclo-stationary, which needs to be incorporated in thereceiver structure as well.

The received signal r at time t for a single user, after propagationthough the wireless channel, is given by: $\begin{matrix}{{r(t)} = {{\int_{- \infty}^{+ \infty}{{h\left( {t,\tau} \right)}{s\left( {t - \tau - \zeta} \right)}\quad{\mathbb{d}\tau}}} + {\eta(t)}}} & \left( {2a} \right)\end{matrix}$where, in the case of a wide sense stationary uncorrelated scattering(WSSUS) channel, τ is path delay, h (t, τ) is channel function, ξ ispropagation delay, s(t−τ−ξ) is the continuous transmitted signalcorrected for delay, η(t) is thermal noise (modeled as additive whiteGaussian noise (AWGN)) at time t. The propagation delay ξ may beincorporated in the channel model itself, but for the sake of clarityand simplicity, the propagation delay ξ is considered a separatevariable.

Substituting equation (1) for the continuous transmitted signals(t−τ−ξ), the received signal r at time t for a single use isrepresented as: $\begin{matrix}{{r(t)} = {{\sum\limits_{i = {- \infty}}^{+ \infty}\quad{{b\lbrack i\rbrack}{\sum\limits_{n = 1}^{N}\quad{c_{n + {iN}}{\int_{- \infty}^{+ \infty}{{h\left( {t,\tau} \right)}{f\left( {t - \tau - {iT}_{s} - {nT}_{c}} \right)}\quad{\mathbb{d}t}}}}}}} + {\eta(t)}}} & \left( {2b} \right)\end{matrix}$

In a MUD system, the received signal r(t) is a conglomerate ofsuperimposed signals transmitted by the multiple users. When the userindex is represented by k for a total of L users, the total receivedsignal at the receiver in the MUD system is derived from equation (2b)as: $\begin{matrix}{{r(t)} = {\sum\limits_{k = 1}^{L}\quad{r^{(k)}(t)}}} & \left( {3a} \right) \\{{r(t)} = {{\sum\limits_{k = 1}^{L}\quad\begin{bmatrix}{\sum\limits_{i = {- \infty}}^{+ \infty}\quad{{b^{(k)}\lbrack i\rbrack}{\sum\limits_{n = 1}^{N}\quad{c_{n + {iN}}^{(k)}{\int_{- \infty}^{+ \infty}{h^{(k)}\left( {t,\tau} \right)}}}}}} \\{f\left( {t - \tau - {iT}_{s} - {nT}_{c}} \right)\quad{\mathbb{d}t}}\end{bmatrix}} + {\eta(t)}}} & \left( {3b} \right)\end{matrix}$where it is assumed that the pulse shaping filter 307 is the same forall the users.

At the receiver, the analog signal is converted to a digital signalthrough a match filter (matched to the pulse shaping filter 307 at thetransmitter). In the case of the downlink model for CDMA systems, suchas IS-95, cdma2000 and W-CDMA, all of the received signals (from thesame base station) may be modeled with the same channel function h andsame propagation delay ξ. In the uplink, however, the different signalsare treated in an asynchronous manner. In other words, each channelfunction h and propagation delay ξ are different for each of thedifferent users.

Assuming that k=1 represents the signal from the desired user, each ofthe remaining signals (k≠1) are treated as interfering signals. Thereceiver is therefore time synchronized to k=1, which implies that thediscrete representation of the received signal r_(n) is given by:$\begin{matrix}{{r_{n}\lbrack i\rbrack} = {\int_{{iT}_{s} + {nT}_{c}}^{{iT}_{s} + {{({n + 1})}T_{c}}}{{r(t)}{f^{*}\left( {t - \xi_{1} - {nT}_{c} - {iT}_{s}} \right)}\quad{\mathbb{d}t}}}} & (4)\end{matrix}$where n is the chip index and i is the bit index. Incorporating themodel for the received signal r(t) as represented in equation (3), thediscrete signal at the receiver is written as: $\begin{matrix}{{r_{a}\lbrack i\rbrack} = {{\sum\limits_{k = 1}^{L}\quad\left\lbrack {\sum\limits_{i^{\prime} = {- \infty}}^{i}\quad{{b^{(k)}\left\lbrack i^{\prime} \right\rbrack}{p_{n}^{(k)}\left\lbrack {i - i^{\prime}} \right\rbrack}}} \right\rbrack} + {\eta_{n}\lbrack i\rbrack}}} & (5)\end{matrix}$where i′ is the summation index and p_(n) ^((k)) is the signature of theuser k, such that p_(k)[(i−i′), n] is the desired signal at thereceiver, representing the received energy in the n^(th) chip of thei^(th) bit from the i′^(th) bit, and is given by: $\begin{matrix}{{{p_{n}^{(k)}\left\lbrack {i - i^{\prime}} \right\rbrack} = {\sum\limits_{n^{\prime} = 1}^{N}\quad{{c_{n}^{(k)}\left\lbrack i^{\prime} \right\rbrack}{\int_{0}^{T_{c}}{\begin{bmatrix}{\int_{- \infty}^{+ \infty}{{h^{(k)}\left( {t,\tau} \right)}{f\quad\left( {t - \tau - \left( {\zeta_{k} - \zeta_{l}} \right) -} \right.}}} \\{\left. {{\left( {i^{\prime} - i} \right)T_{s}} - {\left( {n^{\prime} - n} \right)T_{c}}} \right){f^{*}(t)}{\mathbb{d}\tau}}\end{bmatrix}{\mathbb{d}t}}}}}}\quad} & (6)\end{matrix}$

Assuming that the processing gain N is sufficiently large, i.e.,T_(b)>>T_(c), then p_(k)[(i−i′), n]=0 when i≠i′. Accordingly, asindicated by equation (5), the representation of the discrete receivedsignal in vector notation is given by: $\begin{matrix}{{r\lbrack i\rbrack} = {{\sum\limits_{k = 1}^{L}\quad{{b^{(k)}\lbrack i\rbrack}{p^{(k)}\lbrack i\rbrack}}} + {\eta\lbrack i\rbrack}}} & (7)\end{matrix}$where vector r[i] is a [1×N] dimensional vector representing the signalreceived at the chip level from the k users during the i^(th) bit.Vector p^((k))[i] is a [1×N] dimensional vector representing thesignature of each user k during the i^(th) bit and vector η[i] is a[1×N] dimensional vector representing the thermal noise corresponding tothe received signal r[i]. The i^(th) bit received from user k isrepresented by b^((k))[i].

As previously stated, the user index k increases incrementally from 1 toL users, indicating that, for any one user, there are L−1 interferingsignals. Without loss of generality, k=1 is assumed to indicate thedesired user whose data bit is to be detected. The remaining L−1sources, represented by k≠1, are considered as interfering signals(e.g., MAI). However, not all of the L−1 interfering signals necessarilyrepresent different users. For example, the (i−1)^(th) bit of thedesired k=1 signal may likewise be modeled as an interfering signal(e.g., IAI).

In a conventional single user CDMA detector, incorporating Rakearchitecture, a received composite signal is correlated with theappropriate matched filter function pertinent to the user k to estimatethe value of the corresponding transmitted data bit. The relationshipbetween the estimated detected bit b and the vectors for the signaturesand corresponding received signals is represented as:{circumflex over (b)} ^((k)) [i]=(p ^((k)))^(H) r[i]  (8)where {circumflex over (b)}^((k))[i] is the estimated data bit for userk, (p^((k))) is the Hermitian conjugate of the signature vector p^((k))associated with the user k, and r[i] is the received signal vector forthe i^(th) bit. In addition, estimation of the channel function in aconventional CDMA detector is performed through the pilot channel, basedon a predefined chip sequence (usually un-modulated). However, this CDMAarchitecture tends to be overly complicated for a MUD because thecomputational complexity of the channel estimation and detection of thecorresponding user signals increases exponentially as the number ofusers increases.

In CDMA systems, when the spreading code s is the same for each databit, the chip sequence c[0038] is independent of i. Each of thesignatures p is therefore received only on the channel parameters thatvary from one bit to the next. Although embodiments of the inventioninclude different channel estimation schemes, for purposes ofsimplifying the explanation, a stationary channel model is assumed. Forexample, in W-CDMA, a long scrambling code (e.g., repeating every 10msec. frame) may be replaced with a short scrambling code (e.g.,repeating once every 0.2604 μsec. symbol). Furthermore, representationof the discrete time signal at the receiver assumes that the signal issampled only once per chip. However, alternative detector structures forCDMA may sample the signal at alternative rates, including a rate higherthan the chip duration, i.e., multiple samples per chip.

The present invention applies the foregoing principles of CDMA detectionto an MMSE receiver. An MMSE receiver incorporates use of a dataestimation vector D, such that the mean square error(|b⁽¹⁾[i]−D^(H)r[i]²) of the received signal is minimized, assuming thatuser 1 is the desired user and that b⁽¹⁾[i] is the signal to bedetected. The estimation vector D that achieves the mean square errorminimization is represented by: $\begin{matrix}{D = {{\sigma_{1}^{2}\left( {{\sum\limits_{k = 1}^{L}\quad{\sigma_{k}^{2}{p^{(k)}\left( p^{(k)} \right)}^{H}}} + \Gamma} \right)}^{- 1}p^{(1)}}} & (9)\end{matrix}$where Γ is the covariance matrix of the AWGN, σ_(k) is the averageenergy per data bit k, and σ₁ is the average energy for the data bitk=1. The average energy per data bit is a function of a modulationtechnique. The variables σ_(k) and C₁ also represent the variance of thek^(th) and the 1^(st) data bits, respectively. Equation (9) is derivedas explained in Appendix A, the contents of which are expresslyincorporated herein by reference in their entireties. In particular,equation (9) matches equation (39) of Appendix A.

At a first glance, it appears from equation (9) that, in addition to thesignature of the desired user signal (k=1), the MMSE receiver requiresthe signatures of interfering signals (k≠1) and the covariance matrix ofthe AWGN, in order to determine the optimum diction vector c_(k), forthe k^(th) user, discussed below. However, this is not the case becausethe signals received by the MMUSE receiver inherently includestatistics, by which the interfering signal and the covariance matrixinformation may be sufficiently estimated, without determining theactual parameters. Consequently, the MMSE receiver need only know thesignature p of the desired user (k=1).

The estimating process is accomplished, for example, considering theaverage of r[i]r^(H)[i] taken over several bits, where r[i] is thereceived signal vector for i bits and r^(H)[i] is the Hermitianconjugate of the vector r[i]. Typically, the average of r[i]r^(H)[i] istaken over an entire slot (e.g., 667 μsec.), which is a small enoughperiod of time for the MMSE receiver to assume that the channel isstationary. Therefore, the expectation value of r[i]r^(H)[i] is shownas: $\begin{matrix}{{E\left\{ {{r\lbrack i\rbrack}{r^{H}\lbrack i\rbrack}} \right\}} = \begin{matrix}{{\sum\limits_{j \neq k}^{L}\quad{p^{(k)}p^{{(j)}H}E\left\{ {b^{(k)}\lbrack i\rbrack} \right\} E\left\{ {b^{(j)}*\lbrack i\rbrack} \right\}}} +} \\{{\sum\limits_{j = 1}^{L}\quad{p^{(j)}p^{{(j)}H}E\left\{ {{b^{(j)}\lbrack i\rbrack}}^{2} \right\}}} +} \\{{\sum\limits_{j = 1}^{L}{E\left\{ {b^{(i)}\lbrack i\rbrack} \right\} p^{(j)}E\left\{ {\eta^{H}\lbrack i\rbrack} \right\}}} +} \\{{\sum\limits_{j = 1}^{L}\quad{E\left\{ {b^{(j)}*\lbrack i\rbrack} \right\} E\left\{ {\eta\lbrack i\rbrack} \right\} p^{{(j)}H}}} +} \\{E\left\{ {{\eta\lbrack i\rbrack}{\eta^{H}\lbrack i\rbrack}} \right\}}\end{matrix}} & (10)\end{matrix}$where it is assumed that each of the signatures corresponding to theusers do not depend on the bit index i, which is true for any DS-CDMAsystem using short scrambling codes or PN sequences (e.g., truncatedGold codes). Additional statistical properties of the data bits areassumed, including, for example, that data bits belonging to users otherthan the desired user are uncorrelated and that the average value of thedata bits E{b^((j))[i]} is zero (for all j).

Under the assumption that the bit streams from different users areindependent of each other, the expectation value of r[i]r^(H)[i] isfurther simplified as: $\begin{matrix}{{E\left\{ {{r\lbrack i\rbrack}{r^{H}\lbrack i\rbrack}} \right\}} = {{\sum\limits_{j = 1}^{L}\quad{\sigma_{j}^{2}p^{(j)}p^{{(j)}H}}} + \Gamma}} & \left( {11a} \right)\end{matrix}$Substituting the relationship of equation (11a) into equation (9), thedata estimation vector D may be represented as:D=σ ² ₁ [E{r[i]r ^(H) [i]}] ⁻¹ p ⁽¹⁾  (11b)Therefore, as stated above, the MMSE receiver for the k^(th) user (e.g.,k=1) is required to know only the signature of the k^(th) user (e.g,p⁽¹⁾), and the remaining statistics are derived by taking the average ofthe received signal r[i]r^(H)[i] over multiple bits. In an alternativeembodiment, the averaging is performed over a training sequence, as wellas over the actual data sequence.

As stated above, a significant drawback of conventional adaptive MMSEreceivers is the need for a training sequence. In particular, a trainingsequence is a series of known bits transmitted by the transmitter andused by the receiver to estimate the channel h(τ, t). The estimatedchannel h(τ, t) relates the chip sequence of the desired signal c_(k)(at the transmitter) and the signature of the desired signal p_(k)[i′−i,n] at the receiver, as shown, for example in equation (6). Generally,the training sequence is a time multiplexed signal and thereforeindependent of the data signals.

The adaptive MMSE receiver 200 of the present invention does not use atraining signal. Instead, a common pilot channel and/or a dedicatedpilot channel are used to estimate the channel, similar to those used inCDMA systems. According to the present invention, the channel estimateis used to determine the signature of the desired signal.

One drawback of relying on code multiplexed pilot signals is that acommon pilot channel, for example, introduces residual correlationbetween the desired signal and the pilot itself, due to the orthogonalrelationship between the signals. In comparison, a typical timemultiplexed training sequence has no orthogonality with respect to therelated data signals because the training sequence is transmitted at aseparate time. The residual correlation may cause some degradation inreceiver performance, but the advantages of eliminating the need for atraining sequence outweighs the negative effects. Furthermore, the MMSEreceiver 200 of the present invention does not require significantchanges in the W-CDMA protocol (physical layer or higher layers). Onlyminor changes relating to the pilot signal bit sequences are required,discussed below. Therefore, implementation in a W-CDMA system is greatlysimplified, as compared to a training sequence-based MMSE receiver.

In an embodiment of the invention, the channel vector H of the adaptiveMMSE receiver 200 is related to the spreading sequence (or chipsequence) c^(i) and the signature p^(i) of the i^(th) user as:$\begin{matrix}{{\begin{bmatrix}p_{0}^{i} \\p_{1}^{i} \\p_{2}^{i} \\\cdots \\p_{N - 1}^{i} \\p_{N}^{i} \\p_{N + 1}^{i} \\\cdots \\p_{N + M - 3}^{i} \\p_{N + M - 2}^{i} \\p_{N + M - 1}^{i}\end{bmatrix}\begin{bmatrix}h_{0} & 0 & 0 & \cdots & \quad & \quad & \quad \\h_{1} & h_{0} & 0 & \cdots & \quad & \quad & \quad \\h_{2} & h_{1} & h_{0} & \cdots & \quad & \quad & \quad \\\cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots \\\quad & \quad & \quad & \cdots & h_{2} & h_{1} & h_{0} \\\quad & \quad & \quad & \cdots & h_{3} & h_{2} & h_{1} \\\quad & \quad & \quad & \cdots & h_{4} & h_{3} & h_{2} \\\cdots & \cdots & \cdots & \cdots & \cdots & \cdots & \cdots \\\quad & \quad & \quad & \cdots & h_{M} & h_{M - 1} & h_{M - 2} \\\quad & \quad & \quad & \cdots & 0 & h_{M} & h_{M - 1} \\\quad & \quad & \quad & \cdots & 0 & 0 & h_{m}\end{bmatrix}} \times \begin{bmatrix}c_{0}^{i} \\c_{1}^{i} \\c_{2}^{i} \\\cdots \\c_{N - 1}^{i}\end{bmatrix}} & (12)\end{matrix}$where N is the bit duration and M is the delay spread, in units ofchips, between received bits. As indicated, the signature vector p ismerely the channel's response to the spreading sequence vector c of thereceived data bits. Due to the delay spread of M chips, the signaturevector p of each user persists well into the next data bit. In otherwords, the signature vector p is a [1×(N+M)] vector and the spreadingsequence c is a [1×N] vector.

In an embodiment of the invention, it is assumed that the delay spread Mis less than the bit duration N (i.e., the delay spread M is less thanN). The received signal during the i^(th) bit period is denoted asr[i]d. Therefore, for the n^(th) chip (n=N) during the i^(th) symbolperiod, the received signal is represented as: $\begin{matrix}{{r_{n}\lbrack i\rbrack} = {{\sum\limits_{j = 0}^{1}\quad\begin{bmatrix}{{{b^{pilot}\left\lbrack {i - j} \right\rbrack}p_{n + {jN}}^{({pilot})}} +} \\{{{b^{1}\left\lbrack {i - j} \right\rbrack}p_{n + {jN}}^{(l)}} + {\sum\limits_{k = 2}^{L}\quad{{b^{k}\left\lbrack {i - j} \right\rbrack}p_{n + {jN}}^{(k)}}}}\end{bmatrix}} + {\eta_{n}\lbrack i\rbrack}}} & (13)\end{matrix}$where the bit index j goes from 0 to 1, because it is assumed that thedelay spread M is less that a bit. In alternative embodiments, the sameconcept can be extended to a delay spread M larger than the bit durationN.

The signature of the pilot p^((pilot)) in equation (13) may berepresented, in terms of the estimated channel {h_(k); k=0, . . . M} andthe spreading sequence of the pilot c^((pilot)), such that p^((pilot))_(n+jN)=h_(m)c^((pilot)) _(n+jN−m). Likewise, the signature of the datafor each user p^((k)) (including p⁽¹⁾) may be represented as p^((k))_(n+jN)=h_(m)c_(n+jN−m). Accordingly, equation (13) is may be writtenas: $\begin{matrix}{{r_{n}\lbrack i\rbrack} = {{\sum\limits_{j = 0}^{1}\quad\begin{bmatrix}{{\sum\limits_{m = 0}^{M}\quad{{b^{pilot}\left\lbrack {i - j} \right\rbrack}h_{m}c_{n + {jN} - m}^{({pilot})}}} +} \\{{\sum\limits_{m = 0}^{M}\quad{{b^{1}\left\lbrack {i - j} \right\rbrack}h_{m}c_{n + {jN} - m}^{(l)}}} +} \\{\sum\limits_{k = 2}^{L}\quad{\sum\limits_{m = 0}^{M}\quad{{b^{k}\left\lbrack {i - j} \right\rbrack}h_{m}c_{n + {jN} - m}^{(k)}}}}\end{bmatrix}} + {\eta_{n}\lbrack i\rbrack}}} & (14)\end{matrix}$

The spreading sequences for the pilot and the data signals are [1×N]vectors produced by the modulo 2 addition of OVSF codes of length N andPN sequence. In an embodiment of the invention, the PN sequence (orscrambling code) is based on the short Gold codes. The Gold code isrepeated for each bit, keeping the spreading sequence of each bitstationary.

The OVSF code is used for orthogonalization of the different users (k=1,. . . L) within the system. The purpose of the PN sequences is torandomize the signature of each user and differentiate one transmitter(e.g., Node B base station) from the others in the network. The OVSFcode for the pilot channel is assumed to be all zeros, without the lossof generality, although any other OVSF code that is orthogonal to theother codes suffices.

As discussed in regard to equation (11), for the MMSE estimation to beevaluated without knowledge of the signatures of other users, the databits of all the users, as well as the pilot, must satisfy thestatistical property: E{b^((pilot))[i]}=E{b^((k))[i]}=0. The property issatisfied for the data bits because the data bits are modeled as randomsequences. However, the pilot bits cannot be un-modulated. Becauseconventional W-CDMA systems currently recognize un-modulated pilotsignals, the W-CDMA physical layer must be modified slightly. Inparticular, the W-CDMA protocol is changed to ensure that the pilotbits, b^((pilot))[i], have an expectation value E{b^((pilot))[i]}=0,which is obtained in response to certain predetermined bit sequences,such as {−1, 1, −1, 1, −1, 1, . . . } for binary phase shift key (BPSK)modulation and {1+i, −1+i, −1−i, −1+i, . . . 1+i, −1−i, −1−i, −1+i} forquaternary phase shift key (QPSK) modulation.

As discussed above, the adaptive MMSE receiver 200 of the presentinvention follows a joint channel estimation and signal detectionscheme, requiring a coarse timing estimate and short scrambling codes(i.e., the same from one bit to the next, as long as the channel isinvariant). The channel estimation and the detection of the desiredsignal are performed jointly. In particular, the signal is estimatedbased on the relative phase of the data bits to the pilot bits, at thesame time the channel is estimated based on the pilot bits. Similar toconventional CDMA Rake receivers, the pilot signal in the presentinvention provides the phase reference.

Referring to equation (14), the signal received during the n^(th) chipin the i^(th) bit of the transmitted signal is represented as:$\begin{matrix}{{r_{n}\lbrack i\rbrack} = {{\sum\limits_{j = 0}^{1}\quad\begin{bmatrix}{{\sum\limits_{m = 0}^{M}\quad{{\alpha_{m}\left\lbrack {i - j} \right\rbrack}c_{n + {jN} - m}^{({pilot})}}} +} \\{{\sum\limits_{m = 0}^{M}\quad{{\beta_{m}\left\lbrack {i - j} \right\rbrack}c_{n + {jN} - m}^{(l)}}} +} \\{\sum\limits_{k = 2}^{L}\quad{\sum\limits_{m = 0}^{M}\quad{{\delta_{m}^{k}\left\lbrack {i - j} \right\rbrack}c_{n + {jN} - m}^{(k)}}}}\end{bmatrix}} + {\eta_{n}\lbrack i\rbrack}}} & (15)\end{matrix}$where α_(m)[i−j] relates to the unknown pilot bit (or pilot symbol)b^((pilot))[i−j] h_(m); β_(m)[i−j] relates to the unknown data bit (ordata symbol) of the desired user b⁽¹⁾[i−j] h_(m); and δ_(m) ^(k)[i−j]relates to the unknown data bit of other users b^((k))[i−j] h_(m). The2(M+1) α_(m)[i] variables and the 2(M+1) β_(m)[i] variables are thedesired signals (pilot and data, respectively). The remaining α and βvariables, as well as all of the δ variables, are treated as interferingsignals. The interfering signals include data bits of other users (e.g.,MAI) and data and pilot bits of the same (desired) user that do notbelong to the i^(th) bit (e.g., ISI).

The spreading sequences of the 2(M+1) signals, represented byc^((pilot)) _(n+jN−m)|_(j=0) and c⁽¹⁾ _(n+jN−m)|_(j=0), are determinedbased on the scrambling code (or the PN sequence) and the spreading codes of the transmitter. As stated above, the variables α_(m)[i−j] andβ_(m)[i−j] for j≠0 are considered interference. Therefore, thecorresponding spreading sequences are defined as: $\begin{matrix}{{c_{n + {jN} - m}^{({pilot})}❘_{j = 0}} = \begin{matrix}{{PN}_{n - m}s_{n - m}^{({pilot})}} & {{{if}\quad 0} \leq {n - m} \leq {N - 1}} \\0 & {otherwise}\end{matrix}} & \left( {16a} \right) \\{{c_{n + {jN} - m}^{(1)}❘_{j = 0}} = \begin{matrix}{{PN}_{n - m}s_{n - m}^{(1)}} & {{{if}\quad 0} \leq {n - m} \leq {N - 1}} \\0 & {otherwise}\end{matrix}} & \left( {16b} \right)\end{matrix}$where PN_(i) is the scrambling code, s_(i) ^((pilot)) is the OVSFspreading code of the pilot bit and s_(i) ⁽¹⁾ is the OVSF spreading codeof the data bit. In the depicted embodiment of the invention, the PNsequence is a short Gold code, for example. In alternative embodiments,the receiver may determine and store all spreading sequences of thepilot and data signals based on the initially received signals, orincrementally calculate spreading sequences upon receipt of eachreceived pilot and data signal, in real-time.

The adaptive MMSE receiver 200 collects consecutive signals to populateestimation vectors P_(m) and D_(m), representing the pilot signals andthe data signals, respectively. The number of signal samples collectedfor estimating the vectors depends on the number of chips representingeach bit within the signals, discussed, for example, with respect toFIG. 4, below. Based on the relationship indicated by equation (11b),the pilot and data estimation vectors P_(m) and D_(m) are:(P _(m))_(n) =[E{r[i]r ^(H) [i]}] ⁻¹ c ^((pilot)) _(n−m) for n=0, 1, . .. N+M  (17a)(D _(m))_(n) =[E{r[i]r ^(H) [i]}] ⁻¹ c ^((pilot)) _(n−m) for n=0, 1, . .. N+M  (17b)where r[i]=(r₀[i], r₁[i], . . . r_(N+M−1)[i])^(T), and r^(H)[i] is theHermitian conjugate of the receive vector r[i].

As shown in FIG. 4, the exemplary received signal vector r[i] isconstructed from the N+M chip samples, starting with the beginning ofthe i^(th) bit duration T_(b). T_(b) is the same for the bits i−1, i andi+1. T_(c) represents the chip duration that is equal to 1/10^(th) ofT_(b) (i.e. T_(b)=10T_(c)). The delay spread is four chips, or M=4.Therefore, in the example represented by FIG. 4, each of the pilot anddata estimation vectors P_(m) and D_(m) are calculated from receivevectors based on 14 chip samples.

For a WSSUS channel model, the different multipaths corresponding to thedifferent chip delays (m) are considered independent in terms of fading,enabling a corresponding assumption that the 2(M+1) unknown signalsα_(m)[i] and β_(m)[i] are likewise independent. An MMSE estimationalgorithm is therefore used to estimate the unknown signals α_(m)[i] andβ_(m)[i] based on the received signal vector r[i], the pilot estimationvector P_(m) and the data estimation vector D_(m), according to thefollowing relationships, derived from equation (9), above:$\begin{matrix}{{\frac{1}{\sigma_{\alpha_{m}}^{2}}{{\hat{\alpha}}_{m}\lbrack i\rbrack}} = {{P_{m}^{H} \cdot {r\lbrack i\rbrack}} = {\sum\limits_{n = 0}^{N + M - 1}\quad{\left( P_{m}^{H} \right)_{n}{r_{m}\lbrack i\rbrack}}}}} & \left( {18a} \right) \\{{\frac{1}{\sigma_{\beta_{m}}^{2}}{{\hat{\beta}}_{m}\lbrack i\rbrack}} = {{D_{m}^{H} \cdot {r\lbrack i\rbrack}} = {\sum\limits_{n = 0}^{N + M - 1}\quad{\left( D_{m}^{H} \right)_{n}{r_{m}\lbrack i\rbrack}}}}} & \quad\end{matrix}$Therefore, the estimated values of α_(m)[i] and β_(m)[i] may be shownas:{circumflex over (α)}_(m) [i]=P ^(H) _(m) ·r[i]·σ ² _(α) _(m){circumflex over (β)}_(m) [i]=D ^(H) _(m) ·r[i]·σ ² _(β) _(m)   (18b)

As discussed with respect to equation (15), the variablesα_(m)[i]=(b^((pilot)) [i])h_(m) and β_(m)[i]=(b⁽¹⁾[i])h_(m), whereb^((pilot))[i] and h_(m), b⁽¹⁾[i] are zero mean random variables. Underthis assumption, the variance σ² _(α) _(m) and σ² _(β) _(m)corresponding to α_(m)[i] and β_(m)[i] respectively, can be representedas:σ² _(α) _(m) =σ² _(pilot)σ² _(H) _(m)σ² _(β) _(m) =σ² ₁σ² _(H) _(m)   (19)where σ_(pilot)² = E{b^((pilot))[i]²}    and  σ₁² = E{b⁽¹⁾[i]²}.

In accordance with the definitions of α_(m) and β_(m) in relation toequation (15), the estimated desired data bit b⁽¹⁾[i] may be representedas: $\begin{matrix}{{{\hat{b}}^{(1)}\lbrack i\rbrack} = {{b^{({pilot})}\lbrack i\rbrack}{\sum\limits_{m = 0}^{M}\quad\frac{{\hat{\beta}}_{m}\lbrack i\rbrack}{{\hat{\alpha}}_{m}\lbrack i\rbrack}}}} & \left( {20a} \right)\end{matrix}$Substituting the relationships indicated in equations (18b) and (19),the estimated desired data bit b⁽¹⁾[i] is further derived as follows:$\begin{matrix}{{{\hat{b}}^{(1)}\lbrack i\rbrack} = {{b^{({pilot})}\lbrack i\rbrack}{\sum\limits_{m = 0}^{M}\quad\frac{D_{m}^{H}{r\lbrack i\rbrack}\sigma_{\alpha_{m}}^{2}}{P_{m}^{H}{r\lbrack i\rbrack}\sigma_{\beta_{m}}^{2}}}}} & \left( {20b} \right) \\{{{\hat{b}}^{(1)}\lbrack i\rbrack} = {{b^{({pilot})}\lbrack i\rbrack}{\sum\limits_{m = 0}^{M}\quad{\frac{\sigma_{1}^{2}\sigma_{h_{m}}^{2}}{\sigma_{pilot}^{2}\sigma_{h_{m}}^{2}}\frac{D_{m}^{H}{r\lbrack i\rbrack}}{P_{m}^{H}{r\lbrack i\rbrack}}}}}} & \left( {20c} \right) \\{{{\hat{b}}^{(1)}\lbrack i\rbrack} = {{b^{({pilot})}\lbrack i\rbrack}{\sum\limits_{m = 0}^{M}\quad{\frac{\sigma_{1}^{2}}{\sigma_{pilot}^{2}}\frac{D_{m}^{H}}{P_{m}^{H}}}}}} & \left( {20d} \right)\end{matrix}$

As shown in equation (20d), the desired estimated data bit {circumflexover (b)}⁽¹⁾[i] is generally directly related to the pilot bitb^((pilot)). In particular, the pilot bit provides the phase referencefor the detected data bit, as previously described. Furthermore, thedesired estimated data bit {circumflex over (b)}⁽¹⁾[i] is directlyrelated to the square of the variance (or average energy) of the databit σ² ₁ divided by the square of the variance (or the average energy)of the pilot bit σ² _(pilot). Significantly, the σ² ₁ divided by σ²_(pilot) (e.g.,$\left. \frac{\sigma_{1}^{2}}{\sigma_{pilot}^{2}} \right)$is the inverse of the receiver gain, which is a known value, based onthe transmitted signal. Finally, the desired estimated data bit{circumflex over (b)}⁽¹⁾[i] is directly related to the Hermitianconjugate of the data estimation vector divided by the Hermitianconjugate of the pilot estimation vector, each of which are determinedthrough multiple samplings of the received signals and the respectivedata and pilot chip sequences.

FIG. 5 depicts an exemplary flowchart of the detection process in themodified adaptive MMSE receiver 200, in accordance with the presentinvention. The adaptive MMSE receiver 200 receives spread signalsthrough the antenna 201 at step s510. The receipt of the transmittedsignals is continuous, although detection of each data bit is performedbased on a sampling of discrete received signals, where the number ofreceived signals in the sampling is based on a predetermined number ofchips. The received signals include a pilot bit to be used forestimating the receiver channel and providing a phase reference. Thesignal is demodulated from the RF carrier by the multiplexer 204 andconverted to digital data by the A/D converter 206 at step s511.

At step s512, the signal vector r[i] is populated based on thepredetermined sampling of received signals. The vector r[i] is a [1×N]dimensional vector, indicating the values of r for each of the sampledsignals, where N is the bit duration. The Hermitian conjugate r[i] ofthe signal vector r[i] is likewise determined, enabling the update ofthe expectation value E{r[i]r^(H)[i]} at step s514, as indicated atequation 11(b).

The channel estimator 210 and the data bit estimator 212 simultaneouslyprocess the received signal. Steps s516, s520 and s524 are performed bythe channel estimator 210, while steps s518, s520 and s526 are performedby the data bit estimator 212. At step s516, the channel estimator 210determines the spreading sequence of the pilot bit c^(pilot). Inalternative embodiments, the pilot bit may be a common pilot bitdirected to the entire transmitted signal, or a dedicated pilot bitdirected to the desired data bit c¹ at step s518. The spreadingsequences are generally known quantities, which may be determined basedon initial received signal information, for example, and used over theduration of the transmission. The spreading sequences may also becalculated in accordance with equations (16a) and (16b), above.

At steps s520 and s522, the channel estimator 210 determines the averageenergy estimation per pilot bit σ_(pilot) and the data bit detector 210determines the average energy estimation per data bit σ₁, respectively.In an embodiment of the invention, the receiver 200 simply determinesthe ratio of the average energy estimation of the pilot bit to theaverage energy estimation of the data bit to identify the receiver gain.

At step s524, the estimation vector P_(m) is calculated based on theexpectation value E{r[i]r^(H)[i]} and the spreading sequence of thepilot bit c^(pilot). Likewise, at step s526, D_(m) is calculated basedon the expectation value E{r[i]r^(H)[i]} and the spreading sequence ofthe data bit c¹. The calculation of P_(m) and D_(m) are performedaccording to equations (17a) and (17b).

At step s528, the received data bit is detected by estimating the valueof the data bit {circumflex over (b)}⁽¹⁾ based on a phase differencebetween the data bit and the pilot bit b^((pilot)), the average energyestimation per pilot bit σ_(pilot) and data bit σ₁, and the pilot anddata estimation vectors P_(m) and D_(m). In an embodiment of theinvention, detection of the estimated data bit in step s528 is performedin accordance with equation 20(d), where the desired relationshipbetween the average energy estimation per pilot bit σ_(pilot) and theaverage energy estimation per data bit σ₁ is obtained using the inverseof the gain of the receiver 200.

In an alternative embodiment, as depicted in FIG. 2, detection of theestimated data bit in step s528 is performed in accordance with equation20(a). In particular, the values of α and β are respectively determinedby the channel estimator 210 and the data bit estimator 212, inaccordance with equation (18b), for each of the M chips comprising thetransmitted data bit. A total of M+1 dividers, of which only dividers220, 222 and 224 are pictured, respectively divide each β value by eachcorresponding a value, from m=0 through m=M. The output of each of thedividers (e.g., dividers 220, 222 and 224) are added at the summationblock 230 and adjusted for phase difference in reference to the pilotbit b^((pilot)). The resulting output 232 is the estimated detected databit bill.

Significantly, the pilot aided, simultaneous channel estimation and MMSEdetection of the present invention is fairly resilient to Dopplereffects. As described above, the receiver channel is assumed to bestationary for the duration during which E{r[i]r^(H)[i]} is evaluated,which may include several bits. In the exemplary embodiment, the channelhas been set to one slot. The Doppler effect generally prevents thechannel from being modeled in this fashion because a Doppler relatedphase shift e^(j2πΔft) is typically added to each received signalsample. In other words, each received signal sample r_(n)[i] must bemodified to be r_(n)[i] e^(j2πΔf(iTs+nTc)). However, the Doppler effectdoes not alter the estimation process in the present invention becausethe data signal estimate is derived, in part, from phase reference tothe pilot signal. Because both the pilot phase and data phase aremultiplied by e^(j2πΔft), the overall estimate is unaffected by Doppler.

Although the invention has been described with reference to severalexemplary embodiments, it is understood that the words that have beenused are words of description and illustration, rather than words oflimitation. Changes may be made within the purview of the appendedclaims, as presently stated and as amended, without departing from thescope and spirit of the invention in its aspects. Although the inventionhas been described with reference to particular means, materials andembodiments, the invention is not intended to be limited to theparticulars disclosed; rather, the invention extends to all functionallyequivalent structures, methods, and uses such as are within the scope ofthe appended claims.

In accordance with various embodiments of the present invention, themethods described herein are intended for operation as software programsrunning on a computer processor. Dedicated hardware implementationsincluding, but not limited to, application specific integrated circuits,programmable logic arrays and other hardware devices can likewise beconstructed to implement the methods described herein. Furthermore,alternative software implementations including, but not limited to,distributed processing or component/object distributed processing,parallel processing, or virtual machine processing can also beconstructed to implement the methods described herein.

It should also be noted that the software implementations of the presentinvention as described herein are optionally stored on a tangiblestorage medium, such as: a magnetic medium such as a disk or tape; amagneto-optical or optical medium such as a disk; or a solid statemedium such as a memory card or other package that houses one or moreread-only (non-volatile) memories, random access memories, or otherre-writable (volatile) memories. A digital file attachment to email orother self-contained information archive or set of archives isconsidered a distribution medium equivalent to a tangible storagemedium. Accordingly, the invention is considered to include a tangiblestorage medium or distribution medium, as listed herein and includingart-recognized equivalents and successor media, in which the softwareimplementations herein are stored.

Although the present specification describes components and functionsimplemented in the embodiments with reference to particular standardsand protocols, the invention is not limited to such standards andprotocols. Each of the standards for cellular networks (cdma2000, IS-95,W-CDMA) represent examples of the state of the art. Such standards areperiodically superseded by faster or more efficient equivalents havingessentially the same functions. Accordingly, replacement standards andprotocols having the same functions are considered equivalents.

1. A computer readable medium that stores a program for detecting aspread spectrum signal at a receiver, the spread spectrum signalcomprising a plurality of pilot bits and a plurality of data bits, thecomputer readable medium comprising: an estimating code segment forestimating a channel based on at least one pilot bit; and a detectingcode segment for detecting the plurality of data bits simultaneouslywith the estimating code segment estimating the channel.
 2. The computerreadable medium according to claim 1, in which the plurality of databits are spread in accordance with a wideband CDMA (W-CDMA) format. 3.The computer readable medium according to claim 2, in which theplurality of pilot bits are spread in accordance with a modified W-CDMAformat, comprising a predetermined bit sequence of alternating ones andnegative ones.
 4. The computer readable medium according to claim 1, inwhich detecting the plurality of data bits further comprises determininga phase reference with respect to the at least one pilot bit.
 5. Thecomputer readable medium according to claim 1, in which detecting theplurality of data bits is further based on an inverse gain of thereceiver.
 6. The computer readable medium according to claim 1, furthercomprising: a sampling code segment for sampling a predetermined numberof received signals, wherein detecting the plurality of data bits isfurther based on a data bit estimation vector and a pilot bit estimationvector determined though the sampling.
 7. The computer readable mediumaccording to claim 1, in which the at least one pilot bit comprises acommon pilot signal.
 8. An apparatus for detecting a spread spectrumsignal, comprising a plurality of pilot bits and a plurality of databits, the apparatus comprising: a channel estimator for estimating achannel using at least one pilot bit; and a bit estimator for detectingthe plurality of data bits, concurrently with the channel estimatorestimating the channel.
 9. The apparatus according to claim 8, furthercomprising: a receiver for receiving the plurality of pilot bits and theplurality of data bits; and a demodulator for demodulating the pluralityof pilot bits and the plurality of data bits.
 10. The apparatusaccording to claim 9, in which the bit estimator detects the pluralityof data bits further based on an inverse gain of the receiver.
 11. Theapparatus according to claim 8, in which the plurality of data bits arespread in accordance with a W-CDMA format.
 12. The apparatus systemaccording to claim 11, in which the plurality of pilot bits are spreadin accordance with a modified W-CDMA format, comprising a predeterminedbit sequence of alternating ones and negative ones.
 13. The apparatusaccording to claim 8, in which the bit estimator detects the pluralityof data bits further based on a phase reference with respect to the atleast one pilot bit.
 14. The apparatus according to claim 8, in whichthe bit estimator detects the plurality of data bits further based on anexpectation value of the received signal, determined though sampling apredetermined number of received signals.
 15. The apparatus according toclaim 8, in which the at least one pilot bit comprises a common pilotsignal.
 16. A computer readable medium that stores a program fordetecting transmitted spread spectrum data, the computer readable mediumcomprising: an expectation value determining code segment fordetermining a received signal expectation value based on a plurality ofreceived signals; a channel estimating code segment for determining apilot spreading sequence of the received signals and for estimating atleast one pilot portion of the received signals based on the pilotspreading sequence and the expectation value; a bit estimating codesegment for determining a bit spreading sequence of the received signalsand for estimating at least one data portion of the received signalsbased on the bit spreading sequence and the expectation value; and aspread spectrum data estimating code segment for estimating thetransmitted spread spectrum data based on the at least one pilotportion, the at least one data portion, and a phase reference betweenthe at least one data portion and at least one pilot bit of the at leastone pilot portion.
 17. The computer readable medium according to claim16, in which the at least one pilot bit comprises a common pilot signal.18. The computer readable medium according to claim 16, in which the atleast one pilot bit comprises a dedicated pilot signal.
 19. The computerreadable medium according to claim 16, in which the expectation valuedetermining code segment determines the received signal expectationvalue using a covariance matrix.
 20. The computer readable mediumaccording to claim 19, in which the transmitted spread spectrum data isspread in accordance with a CDMA format.